DC/DC Converter Protection

Introduction

As mentioned in the Preface, one of the functions of a DC/DC converter is to protect the application. At the most simple level, this protection consists of matching the load to the primary power supply and stabilising the output voltage against input overvoltages and undervoltages, but a DC/DC converter is also a significant element ensuring system fault protection. For example, output overload limiting and short-circuit protection not only stops the converter from being damaged if the load fails, but also can protect the load from further damage by limiting the output power during a fault condition. In an application with several identical circuits or channels each separately powered by individual DC/DC converters, a fault in one output channel will not affect the other outputs, thus making the system single fault tolerant. Other converter protection features, such as over-temperature shut-down, are primarily designed to safeguard the converter from permanent damage caused by internal component overheating, but a side-effect is also to shut down the application if the ambient temperature gets too high, thus also protecting the components in the application from over-temperature failure.

Adding isolation between input and output breaks ground loops, eliminates source of interference and increases system reliability by protecting the application against transient damage. The elimination of power supply feedback effects is an important facet of DC/DC converter protection. For example, consider a heavy duty DC motor speed controller. The speed controller circuit needs a stable, noise-free supply to smoothly regulate the motor speed. But the high AC ripple drawn by the motor on top of the DC current can create significant voltage transients that could feed back into the speed controller regulation circuit to cause jitter or instability. An isolated DC/DC converter not only delivers a stable low-noise supply to the speed controller circuit, but by breaking the noise feedback loop also protects the motor from unwanted and erratic control signals that could damage the motor and associated drive chain.

However, a DC/DC converter is also constructed from electronic components that are just susceptible to failure if used outside their voltage, current and temperature limits as any other electronic circuit. This chapter investigates protection measures that may be needed to safeguard the converter itself from damage.

Reverse Polarity Protection

DC/DC converter are not protected against reverse polarity connection. Swapping the VIN+ and VIN- terminals will almost certainly cause immediate failure, so care must be taken to ensure that any input connectors or battery connections are polarised. If the primary supply is transformer, then a rectification diode failure could cause a negative- going output that would then also cause the DC/DC converters to fail.

The main reason why DC/DC Converters fail if reverse polarised is the body diode in the FET. This substrate diode conducts when reverse connected and allows a very large current IR to flow, which can lead to the destruction of components on the primary side. To avoid this potential danger, several options are available.



Series Diode Reverse Polarity Protection

The easiest way to protect a DC/DC converter from reverse connection damage is to add a series diode. Fig. 4.2 shows the circuit. If the supply voltage is reversed, the diode D1 blocks the negative current flow and no fault current can flow through the input circuit of the DC/DC converter. Obviously, by replacing the diode with a bridge rectifier, then the converter will function irrespectively of the input voltage polarity.



The series diode protection has a disadvantage, especially at low input voltages, due to the voltage drop across the diode. Depending on the choice of diode, a forward voltage drop of 0.2V to 0.7V can be expected, with an associated power loss = VF × IIN, which reduces both the conversion efficiency and the usable input voltage range. If the input current is 1A, then a standard power diode with VF = 0.5V dissipates 0.5W, equal to about a quarter of the dissipated power of a typical 15W converter, thus reducing the overall efficiency by 20%.

In some applications, the voltage drop across the diode is an advantage. Rally cars often use a 16V battery to increase the brightness of the headlamps. The alternator is modified to deliver 11 - 20V, outside the range of a standard 9 - 18V DC/DC converter. By using three diodes in series, the effective input range can be dropped to match the standard 18V input voltage range.

Shunt Diode Reverse Polarity Protection

An alternative to the series diode is the shunt diode reverse polarity protection. The forward voltage drop across the diode is avoided, but the primary supply must either be overload protected or a series fuse must be fitted (Fig. 4.3). Although this arrangement might seem at first sight to be a better solution than the series diode form of protection, in practice it has several disadvantages. One disadvantage is that although the voltage across the converter when reverse polarity connected is theoretically limited to -0.7V, even this low level of negative voltage can be sufficient to damage some converters and depending on the source impedance and diode characteristics the voltage can be even higher. Secondly, the choice of fuse is not a trivial task (see section 4.3) and its effect on performance is often underestimated. A fuse is, in effect, a resistor that is designed to burn out at a certain current. As with all resistors, there will be a volt drop across it that is current dependent. A fuse may have an insertion loss similar or even higher than the forward drop of a diode (see table 4.1).



P-FET Reverse Polarity Protection

A third option for reverse polarity protection is to use a series P-FET. The FET is the most expensive solution, but it is still inexpensive in comparison to the cost of the converter. The FET must be a P-channel MOSFET otherwise this solution will not work. The maximum gate-source voltage VGS must exceed the maximum supply voltage or reversed supply voltage. The FET should also have an extremely low RDS,ON resistance, around 50mΩ is an acceptable compromise between component cost and effectiveness. With the supply correctly connected, the FET is biased full on and even with an input current of an amp it will exhibit a volt drop of only 50 millivolts.





To examine the differences between the three different methods of reverse polarity protection, measurements were made using a 12W converter with full load with a worst case 9V input to give a nominal 1.5A input current. As can be seen from Table 4.1, the P-FET solution efficiency is very similar to the circuit with no reverse polarity protection.

Input Fuse

Whether used as an overcurrent protection (failsafe) device without a shunt diode, or used as a reverse polarity protection device with a shunt diode, an input fuse needs to be selected so that it does not blow at the worst case input current during normal operation. As fusewire becomes brittle with age, the fuse rating should be at least 1.6 times the highest input current for a long life. The inrush current during converter start up is significantly higher than the operating current, so the fuse should be of the time- delay type (slow-blow) to avoid nuisance blowing on switch-on. The combination of high fuse current rating and slow reaction time also means that during a reverse polarity fault, the diode must be dimensioned to carry the current and the power supply must also be able to deliver enough current to quickly blow the fuse.

A fuse is a one-time only device. If the power supply is mistakenly cross-connected, then the fuse needs to be replaced before the converter can be used again. This may be an advantage if the circuit should remain permanently disconnected from the supply until the cause of the fault has been eliminated by a maintenance team, but for many other applications it would be preferably to make the application fault tolerant (auto recovery). An alternative to a conventional fuse is to use a resettable protection device, such as a polymeric PTC fuse (PPTC). This is a device similar to a positive temperature coefficient (PTC) resistor that increases its resistance with increasing temperature. Under fault conditions, a PPTC fuse rapidly gets hot until its internal granular structure melts, when it becomes a very high resistance, effectively disconnecting the converter except for a minimum holding current. When the power is removed, the device cools down and automatically resets.

Output Over-Voltage Protection

Over-Voltage Protection (OVP) can be applied to the output or input side of a DC/DC converter. On the output side, the function of the OVP is to protect the application from a regulation fault. Many converters use suppressor diodes as a voltage limiter or “clamp” to ensure that the output voltage does not rise above a certain limit. The difficulty is setting the correct clamp voltage level. A suppressor diode will start to conduct some leakage current well below its trigger point, but setting the trip voltage too high will negate the function of the over-voltage protection. A clamp voltage that is 10% higher than the nominal output voltage is usually an acceptable compromise. Of course, the diode will soon fail if there is a complete regulation failure, as it can only dissipate a limited amount of power, but the clamp is still useful to catch any momentary output spikes that might occur under certain operational conditions.



Input Over-Voltage Protection

The function of OVP applied to the input side of the converter is to make the converter immune to input over-voltage transients or surges and to make the converter conform to EMC regulations and other safety and performance standards.

Due to the growing number of appliances and electronic systems in use today, the frequency of occurrence of electrical glitches on the power supply lines is increasing. There are a number of standards and regulations that define both the amount of power- line conducted interference that an appliance may generate and also how much interference the appliance must withstand (immunity). The immunity tests cover voltage surges, transient and fast transient over-voltages and ESD (Electro-Static Discharge) and are now so arduous, that hardly any piece of electronic equipment can survive the tests without extensive input OVP circuitry.



As all DC/DC converters need a primary power supply, it can be assumed that for most applications that the AC/DC power supply will be fitted with input filters and protection against mains-borne over-voltages, the most severe of which are due to lightning strikes. A typical high voltage surge protector uses a combination of elements such as gas discharge tubes, metal oxide varistors and spark gaps to either divert the energy of the surge to ground or to diffuse the energy over a longer timescale to reduce the peak voltage. Such is the energy contained in a lightning strike, that surge suppressors suffer noticeable degradation with each impulse so they have to be made replaceable.

Therefore DC/DC converters generally do not need to be protected from lightning- induced high voltage surges on the input, with the exception maybe being off-grid powered systems such a photovoltaic power supplies, which will also need lightning strike surge protection. For most applications, there is also no need to offer lightning strike protection on the output side, with the exception maybe being bus-powered systems in industrial plants such as refineries or outdoor lighting systems with long, exposed cable runs. However, as DC/DC converters are directly connected to the primary power supply, they are exposed to the full energy of any over-voltage gliches that do occur and often several layers of overvoltage defence need to be applied.

The following sections cover the principles of OVP protection. The protective measures must always be seen in connection with the source impedance. The lower the source impedance, the more energy the over-voltage spikes contain and the harder and more expensive it is to protect the converter against them. There are two basic protection techniques: crowbar and voltage clamping.

SCR Crowbar Protection

As there are exceptions to the rule that DC/DC converters do not need to be protected from lightning strikes, we will explore one method of DC voltage level protection that can be used to protect the input or output side of a DC/DC converter from very energetic surges, before going on to discuss the more common voltage clamping circuits used to protect the input side from regular spikes and over-voltage transient damage.



A crowbar reacts to an overvoltage by short circuiting the lines on which the surge occurs. The most common protection method is the Silicon Controlled Rectifier (SCR) crowbar. An SCR is a thyristor which is ignited when a predetermined voltage is exceeded and then remains in the conducting state until the current through it drops below a holding current limit. The schematic is shown in Fig. 4.7. The Zener diode D_Z1 sets the trigger voltage. Z_i represents the impedance of the long cable.

The advantage of the SCR crowbar on the output side of a hiccup-short circuit protected DC/DC converter is that when the output is short circuited, the current is automatically interrupted by the hiccup circuit and the SCR is reset. The disadvantage of a crowbar on the input side is that the SCR must absorb both the primary power supply short circuit current as well as the overvoltage short circuit current. Therefore it does not reset automatically after being triggered and it is must be used with an input fuse or PPTC device to interrupt the supply to protect both the SCR and the primary power supply from permanent overload. The input side SCR circuit is the same as in Fig. 4.7 except Z_i can be replaced by a fuse.

Clamping Elements

Clamping protection elements are devices whose resistance does not change linearly with applied voltage – at a certain transition point, the current increases exponentially. Unlike SCR’s, clamps need no resetting, so will return to their previous state without the need to interrupt the supply.

Varistor

A varistor is a Voltage Dependent Resistor (VDR) whose resistance varies with the applied voltage. There are various types of varistors, including selenium and silicon carbide, but the most commonly used are metal-oxide varistors (MOV). A MOV is composed of many microscopic spheres of ZnO pressed together and then sintered. At the grain interfaces, junction effects similar to a semiconductor junction arise, so the internal construction of the VDR can be likened to hundreds of back-to-back diodes connected as an array of series and parallel circuits. If the applied voltage is less than the breakdown voltage of the diodes, very little current flows, but if the breakdown voltage is exceeded, an enormous increase in current occurs. Due to the combination of so many diode junctions in series, the breakdown voltage can be made very high – up to several hundred volts. As the diodes are in back-to-back pairs, the effect is symmetrical and a MOV will protect against both positive and negative overvoltages.



The current-voltage relationship shown in Fig. 4.8 follows a power law as shown in Equation 4.1:



Where k is a component-specific constant and α represents the curvature after the knee point. Typical values of α for different protection components are:

α = 35　 Transient Voltage Suppressor Diodes
α = 25　 MOV
α = 8 Selenium cells
α = 4 Silicon Carbide VDR’s

MOVs have a quick reaction time, so can also suppress transients as well as longer- lasting surges, but are not fast enough to suppress ESD overvoltages in the sub microsecond timeframe. Furthermore, they can be damaged by repetitive over-voltage pulses as any inhomogeneities in the internal grain structure cause local heating effects which lead to a gradual degradation in performance (increased leakage current). Multi- Layer MOVs (MLVs) are an attempt to slow down this degradation so that the device can withstand a greater number of internal failures without failing completely, but if the internal power dissipation becomes too high, all MOVs will melt and catastrophically fail. Therefore, a MOV should always be used with an input fuse. The energy rating (in Joules) is the indication of the lifetime expectation of a MOV to repetitive spikes and is an important component selection factor.

Suppression Diode

Unlike the VDR, the protection offered by a suppression diode is provided by a single diode junction, but it has a much larger cross-section for the current path. Suppression diodes are also called Transient Voltage Suppressors (TVS), Silicon Avalanche Diode (SAD) suppressors or by a variety of other trade names. The unipolar V/I characteristic is the same as for a Zener diode (refer to Fig. 4.9), but suppression diodes are engineered to have a much higher peak-to-average power ratio.



As can be seen in Fig. 4.9, a suppressor diode behaves in the first quadrant (top right) as a normal diode in a forward direction and in the third quadrant (bottom left) as a zener diode in the reverse direction. The third quadrant characteristic is defined by three pairs of values; the nominal voltage V_RM (stand-off voltage) at the reverse current I_RM, which indicates the additional burden the supply due to leakage current, the breakdown voltage V_BR at the reverse current I_R, where the characteristic curve starts to knee and small changes in voltage have a large impact on the diode current, and the clamping voltage V_CL, specified at the maximum permissible current I_PP. The suppressor diode should be chosen so that the normal operating voltages come close to, but do not exceed V_BR. In addition, a current limiting resistor may be required so ensure that I_PP is not exceeded.

As a suppressor diode is a unipolar device, it can only react to positive overvoltages. Therefore most TVS packages contain two suppressor diodes placed back-to-back to clamp both positive and negative spikes. The advantage of suppressor diodes over MOVs is that they do not degrade with repetitive spikes and have lower breakdown voltages with more accurate V_BR values, so can protect both low voltage power and signal lines.



OVP Using Several Elements

The suppression characteristics of individual components often do not cover all of the over-voltage protection requirements. Therefore, it may be required to parallel the various elements to obtain the overall desired properties. As shown in the previous sections, varistors or suppression diodes are suitable for OVP in many DC/DC applications, but sometimes to adequately protect the input of a DC/DC converter both are needed in tandem. MOVs have a high current capacity, but also high clamping voltages. On the other hand TVS diodes have very fast switching times (in the order of nanoseconds) and the V_BR can be made lower, but the power rating is also limited. The general rule applies that the faster a protective element reacts, the less power it can handle. For a full OVP, this means that the protection mechanisms must be put in a sequence in such a way that the element is able to process the largest current must also be the first in line. Fig. 4.11 shows a typical arrangement:



Fig. 4.11 shows an OVP network composed of several stages. The series fuse protects against short circuits if the MOV overheats and fails, otherwise the MOV across the input absorbs most of the energy from the input overvoltage surge. In the time it takes for the MOV to react, the input voltage is clamped by the TVS element, with current limiting provided by the series impedance Z_S. Finally, the input capacitor helps to absorb any remaining pulse energy.



If the input spikes are especially energetic, the TVS elements can be paralleled up to share the current. It is not recommended to parallel up the MOV as this would double the chance of a catastrophic failure. It is better to choose a single part with a higher joule rating.

Z_S can be a resistor, in which is good from a cost point of view, but it should be remembered that it is series with the input so must be also rated for the normal operating current and the overall efficiency will be subsequently impaired. A better, but more expensive, choice is a choke with a series resistance in the 100mΩ range.

OVP Standards

The datasheet performance of OVP elements is theoretical and can only be theoretical in nature since, amongst other things, the real-life success of the protection circuit is also dependent on the robustness of the DC/DC converter components as well as the construction of the overall circuit. Even small PCB parasitic inductances and impedances can dramatically affect the result. Therefore a practical test is required to check the OVP in-circuit behaviour and confirm its performance.

As it is impractical to wait for random overvoltage transients and surges that might occur, a number of testing standards have been defined both nationally and internationally. For example, the international standard IEC 61000-4-5 defines a “Surge" as a voltage transient with a rise time of 1.2μs that decays to 50% of its peak value in 50μs, delivered from a high voltage pulse generator with 2Ω source impedance (from input to input) or 12Ω source impedance (from input to ground). The peak voltage of this 1.2/50μs pulse can be chosen between 0.5kV to 4kV, depending on the installation class of the product. Although it is possible to make your own surge tester (the standard gives instructions), it is better to buy a calibrated piece of test equipment with a known performance.



The standard also defines the effects that may occur after such an overvoltage event:





There are similar international standards that define electrical fast transient /burst immunity (e.g, IEC 61000-4-4: 5/50ns waveform, repeating at 5kHz for 15ms or 100kHz for 0.75ms) and Electrostatic Discharge (ESD) voltage levels.

OVP by Disconnection

The choice of testing protocol is heavily dependent on the end-user application and there are other OVP testing standards that are application specific. For example, the EN50155 railway standard requires surge immunity of 140% the nominal input voltage for 1 second. Such a long duration surge voltage cannot be easily clamped without dissipating excessive amounts of power. One solution is to disconnect the input for the duration of the over voltage to protect the DC/DC converter. There are custom controller ICs that available for this task that incorporate the over voltage detection circuitry and FET gate driver that can disconnect the supply voltage in <1μs (Fig. 4.13).



The disconnect method of OVP is not only useful for long duration over-voltage protection, it is also one of the only reliable protection circuits for very low input voltages. A DC/DC converter’s input voltage of, say, 1.2V cannot be easily protected using conventional OVP elements because the temperature coefficient is a significant source of error: either the diodes will start to conduct at the nominal voltage or the clamping voltage will be too high to be useful.

Of course, the disadvantage of disconnect OVP is that the DC/DC converter is deprived of power during the overvoltage event. For short disconnect durations, the input voltage to the DC/DC can be maintained by adding a large capacitor across the input, but for long duration disconnects, a battery-back up or supercap system may be required. The next section explores this solution.

Voltage Dips and Interruptions

In power distribution systems, sudden load increases can cause significant voltage drops. These short-term dips should ideally have no impact on the subsequent power supply components. To protect a DC/DC converter from input voltage dips and interruptions, the usual solution is to store sufficient energy in a capacitor to keep the converter operational during the brown-out or black-out periods. Fig. 4.14 shows a simple circuit.



The circuit consists of a decoupling diode D and one or more capacitors C. The capacitor C is charged in normal operation to the operating voltage V_IN - V_Diode. With an input voltage dip or interruption, the diode blocks the reverse current flow and prevents the capacitor from discharging back into the supply, so that all of the stored energy in the capacitor C is now available to the DC/DC converter. The voltage on the capacitor now starts to decay as it discharges into the DC/DC converter, but it is a complex relationship to calculate because the DC/DC converter is a constant power device, so the input current is inversely proportional to the input voltage.

The energy stored in a capacitor, E_C, is equal to the capacitance, C, multiplied by the square of the voltage, V_C, where V_C is equal to the input voltage V_IN minus the volt drop across the diode, D.



If at t₀ = 0, the input voltage is interrupted, the voltage on the capacitor will begin to decay exponentially according to the capacitor discharge equation:



The charged capacitor can be discharged until time t1. The time t₁ is the time at which the capacitor voltage V_C is equal to the minimum input voltage V_IN,MIN of the DC/DC converter. The remaining energy in the capacitor is then:



The energy required to back up the input voltage over the period t₀ - t₁ is thus:



This energy must supply the necessary input power over the back-up period t_BACK. The required input power can be calculated from the output power and efficiency to give:



This equation can be rearranged to give the required back up capacitance:



These equations tell us that the larger the back-up capacitance, the longer the back-up time, but as large capacitors take up a lot of board space, there are often physical limitations on the size of C. However, the equations also tell us that the stored energy is proportional to V_C², so the wider the input voltage range of the DC/DC converter the better. The DC/DC converter should be selected so that nominal V_IN is close to the maximum input voltage of the converter to get the maximum back-up time. Additionally, a high efficiency or load derating helps.

There are two disadvantages to the simple circuit in Fig. 4.14. The voltage drop across diode D is an additional loss that will reduce the efficiency during normal operation and the high inrush current to charge the large back-up capacitor can be a problem for the primary supply. Both of these problems can be solved by a variation of the disconnect controller shown in Fig. 4.17 that disconnects on an under-voltage instead of an over-voltage and also has a soft-start function to reduce the inrush current.

Inrush Current Limiting

Often too little attention is paid to the issue of inrush current. All DC/DC converters have an internal filter network to reduce their conducted interference. This filter is at least a simple input capacitor and more often an RC or LC low-pass filter or π-Filter. The best filter capacitors have a very low equivalent series resistance (ESR) which means that when power is applied, they present almost a short circuit across the input terminals. An MLCC capacitor can have an ESR of less than 100mΩ. The inrush current IIR is an event that occurs only on start-up but the peak currents can be orders of magnitude higher than the operating input current. As the input capacitors represent almost a dead short, the current is limited only by the impedance of leads (Z_L) and the internal resistance of the power supply (Z_IS).



In addition to the inrush current due to the input filter capacitors, the DC/DC converter is also trying to start up. The transformer is being energised and the load capacitors are also being charged up. All of these energy flows overlap, so it is common to see several inrush current peaks and troughs before the input current stabilises. Fig. 4.16 shows an example for a 2W converter with a normal input current of 80mA but a peak inrush current of nearly 8A. Although such an inrush current seems frighteningly high for a low power converter, it only lasts for 10μs.



In Point-of-Load architectures, many DC/DC converters are connected in parallel to the intermediate bus supply. So there are many low-ESR input filter capacitors in parallel, which can lead to enormously high inrush currents unless appropriate measures are taken.

In complex systems, the converters can be switched on sequentially to keep the inrush currents under control. Otherwise a circuit such as shown in Fig. 4.17 can be used to control the inrush current:



The inrush current limiter functions by shorting out a current limiting resistor R_L only after the input current has stabilised. The field effect transistor Q₁ is an N-channel MOSFET, which is controlled by the R_C network formed by R₁, R₂ and C₁. At power on, C₁ is discharged and holds the gate voltage low, thus Q₁ is OFF. The input capacitance of the DC/DC converter is slowly charged via R_L and the inrush current is reduced. Meanwhile, C₁ charges up via R₁ until the gate of Q₁ reaches the voltage V_IN × R₂/(R₁ + R₂). This voltage is chosen to be sufficient to turn on the FET which than shorts out the series resistor R_L. For small values of C₁, the gate capacitance CG can be significant factor in the timing, but can be calculated using the charging time constant τ = (R₁ || R₂) (C₁ || C_G). The FET should be selected so that it can continuously conduct the worst case input current (maximum output load with a minimum input voltage). R₁ and R₂ should be dimensioned so that the gate voltage is higher than the specified minimum value for VGS at the minimum input voltage.

The choice of RL is up to the user, but usually a few ohms is sufficient to reduce the inrush current to an acceptable level without starving the DC/DC converter of sufficient power to let it start up properly. The ETSI standard ETS 300 132-2 defines the maximum permissible inrush current to be 48× the nominal input current. For the example given in Fig. 4.16, a 6Ω series resistor would be sufficient to make the converter ETSI compliant. In practice, as the converter is low power, the loss through the resistor would only be 40mW during normal operation and the inrush limiting circuit in Fig. 4.17 would be superfluous.

In some applications a NTC can be used as an inrush current limiter. The NTC has initially a high resistance which limits the inrush current. As the device heats up, it reduces its resistance and allows the DC/DC converter current to increase. Its main disadvantage is that the NTC must continuously dissipate power to remain warm enough to maintain its low resistance state.

Load Limiting

Another way to reduce the inrush current is to reduce the load on the converter during start up. This lowers the load dependent part of the inrush current and leaves just the part due to the input filter capacitance. There are two basic ways of load reduction: output soft start and output load switching.

Output soft start works with converters with an output voltage adjust function powering mainly resistive loads only. As the output current is proportional to the output voltage, if the output voltage is initially set low, the output current will also be low and therefore the inrush current will also be low. The output voltage is then allowed to ramp up to the operating voltage.



Fig. 4.18 shows an example of this method using the RECOM R-6112x buck regulator series. When power is applied to the converter, the capacitor C₁ is discharged. The PNP- type transistor is turned on fully and the V_ADJ pin is pulled up to the V_OUT+ voltage. This causes the output voltage to be set to the minimum output voltage. For example, with the R-6112x converter with a nominal 12V @ 1A output, the output voltage will be regulated down to 3.3V at 275mA or about a quarter of full load. As C₁ charges up via R₁, the current through TR1 decreases and the output voltage ramps up to eventually reach the full nominal output voltage. Diode D₁ ensures that C₁ is rapidly discharged when the converter is switched off, ready for the next output soft start.

The second method of inrush current limiting is load switching. This method works with any converter or type of load. The output load is only applied once the output voltage has stabilised. The inrush current therefore has a double peak; once with switch on and once with load on. This spreads the total inrush current over a longer time and reduces the maximum peak current. The output load switcher is a variation on the inrush limiting circuit shown in Fig. 4.19. The field effect transistor Q₁ is an N-channel MOSFET, which is controlled by the R_C network formed by R₁, R₂, R₃ and C₁. At power on, C₁ is discharged and holds the gate voltage low, thus Q₁ is OFF. C₁ then charges up via R₁ until the gate of Q₁ reaches the voltage V_RL × R₂/(R₁ + R₂). This voltage is chosen to be sufficient to turn on the FET and connect the load. R₃ is a high resistance that with capacitor C₁ filters out the resulting output voltage dip due to the load being suddenly switched on. R₁ and R₂ are to be dimensioned so that the gate voltage is higher than the specified minimum value for V_GS at the nominal output voltage. Diode D₁ ensures that when the converter is switched off that C₁ is rapidly discharged, ready for the next switch on cycle.



Under Voltage Lockout

If the input voltage is too low, the input current can exceed the design limits of the DC/DC converter components. Therefore some converters have an internal control circuit that disables the converter if the input voltage drops too low. This circuit is called an Under Voltage Lockout (UVL).

The usefulness of an UVL circuit should not be underestimated. Take, for example, an application that requires 12W of output power with a 12V supply. At nominal input voltage, a 1A supply would suffice, so maybe a 1.5A primary power supply is specified. The converter will typically have a 9 - 18V input voltage range, so at power up it will start to function as soon as the input voltage ramps up above 9V, drawing 1.3A. However, without an UVL circuit, the converter may attempt to start up at only 7V, even though this is out of specification, drawing 1.7A. This is higher than the current limit of the primary supply at the input voltage would collapse. The power supply and converter can interact with each other for several cycles before the converter eventually starts-up properly. In the meantime, the load is presented with several uncontrolled voltage pulses which could damage the application.

Thus a UVL function protects not only the DC/DC converter but also the load and primary power supply from over-current. If the DC/DC converter is not available with a built-in UVL function, the external circuit shown in Fig. 4.20 can be used to disable the converter until the input voltage has stabilised. An op-amp with built-in reference voltage such as the LM10 is a suitable choice.