Understanding Datasheet Parameters

Every respectable manufacturer supplies a technical datasheet with their product that details at the very least the basic operating parameters, overall dimensions and pin connections, but to compare one DC/DC converter with another just relying on the datasheet information often requires interpretation rather than just a simple comparison of numbers.

The problem is that many of the specifications are inter-related so some parameter fixing is needed, i.e, that specific values such as the ambient temperature or input voltage are kept constant during the measurement of the performance specification of interest. For example, a load regulation figure will be made with nominal input voltage, 25°C ambient temperature and be valid over a specified load range. But there are no agreed standards between manufacturers over the parameter fixing, so some will specify a regulation value for the whole 0% - 100% load range, others for 10% - 100% and still others for 20% - 80% load. This means a load regulation specification of ±5% for a load range of 10% - 100% is not necessarily worse than a rival converter with a load regulation specification of ±3% for a load range of 20% - 100%. Similarly a converter with a reliability specification of 1 million hours according to MIL-HDBK-217E is not necessarily more reliable than a converter with a reliability specification of “only” 800 thousand hours according to MIL-HDBK-217F or another converter with “only” 400 thousand hours according to Bellcore/Telcordia.

An unscrupulous manufacturer can use this lack of standardisation to present their product in a better light. A classic example is the output ripple and noise specification, usually given in millivolts peak-to-peak (mVp-p). A converter with 50mVp-p is better than one with 100mVp-p, right? Well not if the fine print at the back of the datasheet states that the first converter measurement was made with a 47μF electrolytic in parallel with a 0.1μF MLCC across the output pins to additionally filter the output and the second converter specification was made without any external components. Additional filter components may be in some cases necessary in order to get a reliable, repeatable measurement, but the customer should be aware that the way the measurement is made affects the measured value and a comparison between two converter specifications can only be done if both are known. In many cases, it is necessary for the customer to measure the critical specifications of concern themselves using the actual or anticipated operating conditions of the application. For example, datasheets do not usually give efficiency vs operating temperature graphs (although RECOM can supply such detailed information on request).

Measurement Methods – DC Characteristics

As already mentioned, the electrical behavior of a DC / DC converter is determined by many different parameters that are specified in the data sheet. To quickly and efficiently characterise a converter and check the validity of the datasheet, it is often worthwhile to use a measurement matrix where the various combinations of load and input voltage can be compared. Table 3.1 shows a typical a measurement matrix setup.





To obtain good and reliable measurement values, the user should take a few basic precautions on how the measurements are made. When preparing the test set up, make sure that the contacts to the DC/DC converter have very low resistance. Often measuring terminals have variable contact resistances, so the best test setup is a "Kelvin" contact, as shown above in Fig. 3.1, where the current and the voltage paths are connected separately to the pins. It is often tempting when using multimeters to stack the 4mm connectors in the meter sockets to connect up two or more meters, but this can lead to significant measurement errors. Each meter should be separately connected to the converter pins as shown above.

To load the DC/DC converter, power resistors or power rheostats can be used, but it is more elegant to use an electronic load. However, some electronic loads need a minimum input voltage to regulate the current properly, so for converter output voltages below 4V, often power resistors are the only choice. A bench power supply makes a good adjustable power supply, but make sure that it can deliver the necessary voltage and current to cover all of the input test requirements. It may be necessary to combine several power supplies to deliver V_IN,MAX. The current limit must be set so that there is sufficient power to supply the DC/DC converter even at the lowest input voltage. Finally check the polarity before turning on – the majority of DC/DC converters are not reverse polarity protected.

Measurement Methods – AC Characteristics

Simply to take an oscilloscope, connect a standard probe to the converter and read the results off the display is not always reliable if the interference mechanisms and their interrelationships are not known. Differential mode (DM) and Common Mode (CM) effects can distort the readings. Section 5 describes DM and CM interference in more detail, but for now, it is sufficient to know that a simple oscilloscope probe largely ignores DM interference as it is symmetrical and occurs on both connections simultaneously, thus the DM component of the AC measurement is missing from the oscilloscope display.

Another source of AC measurement error is the bandwidth capability of the oscilloscope. Oscilloscopes today have an input bandwidth of 400 MHz or more. A closer study of the data sheet, however, reveals that the measurement of output ripple is typically carried out with a bandwidth limit of 20 MHz. This is because on one hand the CM element beyond 20MHz is not so significant because it can be easily filtered out with a small capacitor and on the other hand the measurement should not be dependent on the type or manufacturer of the oscilloscope. An oscilloscope used without the 20MHz BW limitation option will always give higher readings.

Finally, the probe itself can be a source of error. Care must be taken that the cables to the probe are as short as possible. Ideally, the tip of the probe touches to the + pin and the ground pin touches the ring. The use of the supplied earth clip must be absolutely avoided as the loop formed by the earth wire forms an aerial that picks up significant extraneous noise.





If a probe with short contact paths cannot be used, then the proposal shown in Fig. 3.3 is useful. The impedance matching RC components avoid RF reflections that could interfere with the reading.



Note that the measured waveform is halved by the potential divider formed by the two 50 Ohm resistors, so the oscilloscope display should have ×2 multiplication. Even with the matching components, the coax cable should be kept as short as possible.

Measuring Minimum and Maximum Duty Cycle

In some applications, it would be useful to know more about the internal modulation of a DC/DC converter, the duty cycle signal is often not directly accessible from outside the module. However, with some experience, an interpretation of the input or output noise can reveal this information.



The minimum duty cycle δ_MIN is determined by the parameters V_IN = V_IN,MAX and I_OUT = I_OUT,MIN, the maximum duty cycle δ_max by V_IN = V_IN,MIN and I_OUT = I_OUT,MAX. The period T is constant, because it is the operating frequency of the DC/DC converter. Fig. 3.4 shows how the duty cycle can be extracted from the input current waveform.

Output Voltage Accuracy

The Output Voltage Accuracy characteristic, also called the Set Point Accuracy describes the specified tolerance of the output voltage. The parameter is usually specified in percent of the nominal output voltage, typically at room temperature, full load and nominal input voltage.



Output voltage inaccuracy occurs because of component tolerances, especially in the resistor divider that drops the output voltage down to the reference voltage of the PWM comparator (refer back to Fig. 1.46). For output voltages higher than 1.5Vdc, it is common that a 1.22V bandgap voltage reference is used ( a bandgap reference uses two PN junctions arranged so that the temperature coefficient of one junction cancels out that of the other to make a very stable reference voltage). For a 5V output voltage, the resistor divider will have a ratio of 3:1 so if 1% tolerance resistors are used, the output voltage accuracy will be 3%. In addition, the nearest standard value resistor may be used instead of the ideal value, so introducing another error.

Some regulated converters have a trim capability, with which the output voltage can be adjusted within a certain range, typically ±10%. In this case, this specification applies with the trim pin left open (unused).

Unregulated converters have an output voltage that is load dependent. If the nominal output voltage was set to be accurate at 100% load, then the output voltage would be higher than the nominal voltage for all loads below 100%, which could reduce the useable load range of the converter. Therefore, the output voltage is typically set to be accurate at around 60% - 80% load (refer to Fig 1.31). At full load the output voltage is thus always slightly below V_NOM.

Output Voltage Temperature Coefficient

Although the internal bandgap voltage reference has a very stable voltage over a range of operating temperatures, there will be some variation. The resulting temperature coefficient (TC) of the output voltage is defined as the relative deviation of the output voltage at the operating temperature extremes compared with the nominal output voltage at room temperature. The information is usually given in %/C or ppm/K. (ppm = parts per million). Tempcos are typically positive at low temperatures and negative at high temperatures.

To determine the TC, a thermal chamber cabinet is required which can generate the necessary ambient temperatures. At room temperature TRT the nominal voltage V_OUT(T_RT) is measured at nominal load after the DC/DC converter has been run for a 20 minutes waiting time to allow for thermal stabilisation. Likewise the same procedure is carried out at the extremes of the operating temperature range. The calculation of the TC is carried out according to the equation below.



A typical TC value is ±0.02%/C meaning that if the output voltage is nominal at 25°C, the voltage would reduce by 1% at +75°C and increase by 1% at -25°C.

Load Regulation

Load regulation is defined as the maximum deviation of the measured output voltage over the permissible load range from the minimum load (ML) to full load (FL), given as a percentage. The input voltage is typically kept constant at the nominal value, V_NOM. It should be noted that for many converters, a minimum load is required for proper regulation and there may be overload protection, so it is not valid to extrapolate the load regulation figure beyond ML or FL.

Normally, the output voltage changes linearly with the load current, so only two measuring points are needed within the specified load range for the load regulation calculation. If the specifications are done with a nominal current of, say, 80% of the maximum, then the load regulation can be measured in three ways: output voltages at ML and half load (half load = (ML+FL)/2), output voltages at FL and half load, or output voltages at ML and FL, all of which should give approximately the same result. These different calculations are possible from the test setup shown in Fig. 3.1 and the measurement scheme in Table 3.1. The following equation gives the result based on measurements made with ML and FL:



If the datasheet specifies the Output Voltage Accuracy (OVA) figure at 50% load and states a load regulation of ±1%, then at full load, the relative change of the output voltage is -1% and at minimum load the relative change in the output voltage is +1%. Thus the measured output voltage can be higher or lower than the OVA figure by up to 1%. It can be confusing if the Output Voltage Accuracy is specified at full load, as then the load regulation result can only be negative, but by convention a ± percentage is still given. This means if the load regulation is specified at ±1% with OVA specified at 100% load, then the measured voltage can only be equal to or up to 1% lower than the OVA figure. This is in fact twice as accurate as the first definition.

Unregulated converters use Deviation against Load, measured with nominal V_IN to indicate how the output voltage varies with load because they have neither load nor line regulation.

Cross Regulation

This parameter only applies to converters with bipolar or multiple outputs. One output is fixed at full load and the other has a lower load, typically 25% (low load). Then the loads are switched so the first has 25% load and the second output 100% load. The cross regulation in percent is derived from whichever of the two following equations gives the highest figure:



Line Regulation

Line regulation is the deviation of the output voltage due to variations of the input voltage within its minimum (VL) and maximum limits (VH). The load is kept constant, usually at maximum current. Line regulation is specified as the percentage deviation of the output voltage relative to the nominal value of the output voltage. As with load regulation, the output voltage varies linearly with the input voltage, therefore the nominal input voltage (VN) and the difference between VL and VH may be used for determining this parameter. The following equation is based on output voltage deviation measurements with the full input voltage range VL to VH.



The nominal input voltage is usually defined approximately in the center of the input voltage range, so if the datasheet specifies the line regulation to be ±1%, for example, this means a change in the input voltage from VN to VH causes an increase in the output voltage of +1% and a change in the input voltage from VN to VL causes a drop in the output voltage of -1%.

Unregulated regulators are not line regulated. For a fixed load, the output voltage will increase with increasing input voltage and decrease with decreasing input voltage. The relationship is typically not 1:1 though, a 1% change in input voltage does not necessarily cause a corresponding 1% change in the output. The effect of input voltage on output voltage is specified at full load in the format “x%/1% of V_IN”. For example if the line regulation of an unregulated converter is given as 1.2%/1% of V_IN, then the output voltage will increase by 1.2% for every 1% increase in input voltage.

Worst Case Output Voltage Accuracy

The worst case output voltage is the combination of the limits of the output voltage accuracy, load regulation over the load range used, line regulation over the input voltage range used and tempco. As the errors are accumulative, the order in which the calculation is done has an effect, but typically each error can be treated separately to get a first approximation of the output voltage limits:



For example, if the nominal output voltage is 5V, the output voltage accuracy = ±2%, the load regulation ±0.5%, the line regulation ±0.3%, and the TC over the operating temperature range +1.2%/-1.3% then:



Calculating Efficiency

The efficiency of the voltage conversion is given by the ratio of output power to input power. At zero load, the efficiency is always zero. A specification in percent is common, but it can also be given as a normalised number (≤ 1). Normally, the data is provided under several conditions, such as nominal input voltage and full load. To illustrate the complexity of this parameter, Fig. 3.5 shows the efficiency curves of the same DC/DC converter under different measurement conditions.



Input Voltage Range

The input voltage range of DC/DC converter is defined as the input voltage limits VL and VH in between which the converter functions properly with a guaranteed regulated output voltage. Most converters will continue to operate outside of this range, but may not meet all of the datasheet specifications. Higher power converters may have an under voltage lockout (UVL) circuit that shuts down the converter if the input current rises too high due to too low input voltage. This is to protect the converter from input overcurrent damage. Sometimes “Absolute Maximum Ratings” are given in the datasheet to specify the maximum input voltage the converter will accept before the voltage limits on the internal components are exceeded.

The input voltage range given in the datasheet is for continuous voltages. Often higher transient voltages can be applied without causing damage. For example the R-785.0-0.5 buck switching regulator with 5V output has an input voltage range of 6.5V to 32V and an absolute maximum input voltage of 34V, but will withstand a 50V/100ms surge transient and a 1000V/50μs fast transient.

Input Current

The input current is composed of two components, a DC component (the average input current) and an AC component (the back ripple current). The measurement of the back ripple current will be discussed in detail in section 5.2.1. The DC component of the input current is in turn composed of two components, the bias current and the input current due to the load. To find out the bias current, the load can be simply disconnected. The bias current is commonly also referred to as the no load quiescent current (I_Q) or housekeeping current. This current arises because the converter is still oscillating and dissipating power due to the various switching and parasitic losses, and the internal voltage regulators and voltage references are still operating, even though no output current flows. The bias current will be dependent on the input voltage and on the ambient temperature, so I_Q is typically measured at V_IN,NOM and 25°C room temperature. Converters that have an On/Off control or Standby mode can reduce the quiescent current still further by disabling the internal oscillators and regulators as well as the output power stage. Therefore I_OFF is always less than I_Q.

The load-dependent part of the input current is not always easy to interpret. Primarily it depends on the input voltage, so the relationship minimum input voltage = maximum input current is valid, but efficiency against load is non-linear (as shown in Fig. 3.5) and makes the observation difficult. The efficiency is therefore a complex function of the output current and the input voltage. The developer who wants to calculate the maximum input current must know the minimum possible input voltage, the maximum load that could occur in that situation and the efficiency of the converter under those conditions (for example, by reading the efficiency values off a graph like shown in Fig 3.5). It is often inaccurate to make the calculation assuming the full load efficiency given in the datasheets, especially at other loads than full load.

Short Circuit and Overload Current

The output short circuit (S/C) current is the output current that flows when the output pins are connected to each other. A short is typically defined as a connection having a resistance of <1Ω or a low enough shunt resistance that the resulting output voltage is below 100mV. For a single output converter, the short circuit test is between VOUT+ and VOUT-. For a bipolar output converter, the short circuit test can be between VOUT+ and VOUT-, VOUT+ and common or VOUT- and common.

Low power, unregulated DC/DC converters are often not short circuit protected. The industry convention is to claim short circuit capability for 1 second. This is typically the time taken for the internal components to overheat and burn out. So, before a short circuit test is carried out, the developer must first make sure whether the converter is S/C protected and if so, what kind of protection is used: power limiting with thermal shutdown, current foldback or hiccup protection.
Overload protection is not the same as short circuit protection. If the output current exceeds a set limit, typically 110% - 150% of the rated output current, then a current- limited DC/DC converter will allow the output voltage to decrease to keep the current steady at this limit. If the load is increased further, then the output voltage will decrease proportionately. The converter is in constant output power mode instead of constant output voltage. If the overload is removed, the converter goes back to normal operation, but if the overload persists for a long time, the increased internal power dissipation will cause the converter to overheat and either fail or go into thermal shutdown.
However, if the output is short circuited, the output current will still be limited to the set limit, but the output voltage will be very low, theoretically zero for a perfect short, but in practice a few millivolts. The output power is then also close to zero and the converter can operate indefinitely as long as the internal components are rated for the higher current. Thus it is possible for a converter to fail in an overload condition, but survive an indefinite short circuit undamaged. A variation on the current limited protection is current foldback protection (Fig. 3.6).



When the output current exceeds the set limit, the limit is reset to a much lower limit. The DC/DC converter is in power limiting mode, but at a much lower power than in normal operation. This mode usually has to be reset by disconnecting the converter from the supply. While current limiting or current foldback are very effective short circuit protection methods for low to medium power DC/DC converters, they can be ineffective for higher power converters.

If a 1W converter has a 150% current limit, then the converter must cope with 500mW of additional power dissipation during overload or short circuit conditions, but a 50W converter must handle 25W additional power dissipation. This would rapidly overheat the internal components but to over-specify them to carry this high fault-condition current may not be financially justifiable.
The solution to this problem is to use hiccup protection. When the output current exceeds the set limit, the converter immediately shuts down. After a short delay, the converter attempts to restart. If the output current still exceeds the limit, the converter shuts down again and the cycle repeats.
The advantage of hiccup protection is that if the fault condition is removed the converter automatically restarts at the next hiccup. Another advantage of this form of protection is that although the short output pulse causes momentarily high internal power dissipation, the long wait period between hiccups allows the internal components to cool down again, thus the converter runs cold into a direct short circuit.
The disadvantages of hiccup protection are that high capacitive loads can trigger the hiccup mechanism and the converter simply refuses to start up into high capacitive loads. Another disadvantage is if the DC/DC converter is used to supply a bus voltage on long cable networks. A short anywhere on the line will trigger the hiccup mechanism and the hiccup current spikes can make it very difficult to locate the position of the fault.



The easiest way to test the short circuit function with a hiccup-protected DC/DC converter is to simply listen. A converter with hiccup protection makes an audible “tick” each time it attempts to restart. Alternatively an oscilloscope with a current shunt can be used to monitor the output.

To measure current-limited or current fold-back performance, the test setups shown in Fig. 3.8 can be used. In the upper test setup the digital multimeter (DMM) is used in DC current measuring mode and the internal shunt resistor is used as the short circuiting element. This must be monitored to check that the voltage at the DC/DC converter output terminals does not exceed 100mV. For larger short circuit currents, which would exceed the measuring range of the DMM, or cause a larger voltage drop than 0.1V, an external current shunt should be used. The shunt resistor value is selected so that RS < 0.1V/I_SHUNT and PV > 0.1V I_SHUNT.



Remote ON/OFF Control

In many systems it is desirable to be able to turn on and off the DC/DC converter remotely. This can be for reasons of efficiency to reduce the energy consumption or to control the power up and power down sequencing or for safety reasons. Therefore, many DC/DC converters have a control input (on/off control pin) with which the converter can be switched into standby mode. The control pin is an easy pin to drive as any open- collector signal or NPN transistor can be used to control the converter because it only needs a few milliamps of drive current to switch even a high power converter.

Attention should be given to the type of control logic. Positive logic means that the converter is ON with a high or logic ‘1’ signal and OFF with a low or logic ‘0’ signal. The control input is pulled high internally, so if it is left unconnected, the converter is ON. This variant is more commonly used because the converter is active if the control pin is not needed.

Negative logic means that that the converter is OFF with a high or logic ‘1’ signal and ON with a low or logic ‘0’ signal. The control input is pulled high internally, so if it is left unconnected, the converter is OFF. This type of control is useful in a safety critical application where the converter may only be powered up if certain external conditions are first fulfilled.

For an isolated converter, the datasheet should also state to which other pin the on/off control is referenced to. In most cases, the reference potential is the ground of the primary circuit, but in some converters the on/off control is on the output side and is referenced to the secondary VOUT-. If the enable signal originates on the primary side, an isolation element such as optocoupler must then be used to switch the output.

All control pins inputs should have some hysteresis to eliminate repetitive switching with a slowly rising control signal, a situation that can occur, for example, if an external RC delay circuit is used on the control pin to make the converter wait until the input voltage has stabilised before attempting to start up (Fig. 3.9). The datasheet specifies the remote pin voltage, V_REMOTE, as thresholds. Typical values are logic ‘0’ when 0 < V_REMOTE < 1.2V and logic ‘1’ when 3.5 < V_REMOTE < 12V. This means that with a rising V_REMOTE voltage, the converter will switch on when the voltage exceeds 1.2V and with a falling V_REMOTE voltage, the converter will switch off when the voltage drops below 3.5V. The diode in the time delay circuit ensures that the timing capacitor rapidly discharges if the input voltage is switched off so that the time delay remains consistent if the power is then rapidly reapplied.



Isolation Voltage

In isolated DC/DC converters the primary and secondary are separated by the transformer isolation and optocoupler isolation, meaning that there is no direct current path between the two circuits, to give what is known as galvanic separation. The isolation voltage describes the nature of this separation. A high test voltage is specified, either DC voltage or the root mean square value of an alternating voltage, and no significant current should flow when the voltage is applied between the primary and secondary sides.

NOTE: A high potential (HiPot) tester with an accurate current limiting circuit must be used for these kinds of tests as hazardous voltages are used. Do not carry out the HiPot test on an ESD protected worktable as the surface has been treated to make it electrically conductive. Ensure that the HiPot tester has an emergency stop button and make sure that the earth connection to the tester is sound. The Device under Test (DUT) must be adequately insulated from any part that the operator could accidentally touch and the tester should incorporate automatic discharge circuitry to discharge the test voltage after the test has been completed. Follow the maker’s instructions to the letter!

While a converter may be DC galvanically isolated, a leakage current will flow if an AC isolation test is used. The AC current flows through the capacitive coupling between the windings in the transformer and through any EMC suppression capacitors placed across the isolation barrier. Therefore, for an AC HiPot test, not only the RMS voltage but also the permissible leakage current should be specified. Typical set limits are 1mA or 3mA, as any higher leakage currents as this could permanently damage the converter.

Due to the AC leakage current, an AC high voltage stresses the isolation barrier more than an equivalent steady state DC voltage. The stress increases with the frequency and voltage because:



For this reason, a converter that is rated at 1kVdc/1 second should only be tested with 700Vac/1 second (when using a 50Hz signal). This sounds logical if you think that the peak voltage of a 700VacRMS waveform is 980V. However, if the frequency was increased, the leakage current will increase as well. A 100Hz test signal will generate double the leakage current than a 50Hz test signal. So, a converter that is rated at 1kVdc/1 second should only be tested with 360Vac/1 second when using a 100Hz test waveform. Fortunately, a 50Hz HiPot test frequency is the industry standard and although most manufacturers don’t mention the test frequency used, it can be safely assumed that 50Hz was used when comparing isolation values in datasheets. RECOM has a useful isolation voltage comparison tool on it’s website.

The equivalence between DC and AC HiPot testing is also not so straightforward with longer tests than 1 second. A 60 second test puts a lot more strain on the isolation barrier because of the phenomenom known as partial discharge (PD). Partial discharge describes the momentary breakdown due to the high applied voltage between coupling paths within an insulating medium due to internal voids or inconsistencies.

Consider the construction of a conventional enamelled transformer wire. (Fig. 3.10). The insulation lacquer is typically applied in several stages, so there could be discontinuities between layers as well as voids within the insulation.



Momentary current flows as the partial discharges occur within the isolation medium, but the wire is still insulated. However, the voltage stress now has a reduced thickness of insulation barrier. The voltage stress can jump from one weakness to the next until it eventually can cause a complete input/output isolation failure.

The keyword here is “eventually”. It takes time for PD failures to join up and cause a total failure. The longer the HiPot voltage is applied, the more likely that a failure can occur. Thus a HiPot test for 1 minute is much more stressful than the typical production test of 1 second. A converter rated a 1000Vdc/1 second should only be tested at 500Vac/1 min to reduce the likelihood of such cumulative PD failures causing a problem.

Due to the potential of causing permanent damage to the converter during a HiPot test, there are two important practical issues that need special care when setting up a test. Firstly, it is important not to allow voltage gradients to develop within the converter as they can rapidly exceed the breakdown ratings of the internal components. Therefore before carrying out a HiPot test, all of the inputs must be shorted to each other and all of the outputs shorted together. Secondly, as HiPot tests stress the converter isolation and cause cumulative damage to the insulation, it is recommended to reduce the HiPot voltage by 20% for each re-test.



The advantage of HiPot testing is that all of the potential failure paths are checked when a high voltage is placed across the input/output isolation barrier, so the overall dielectric strength of the converter can be guaranteed by a pass result. The disadvantage of the test is that a fail result is destructive – the failed converter must be scrapped.

There is an alternative to HiPot testing, the PD Tester. The test equipment monitors the voltage spikes caused by the partial insulation breakdowns and displays this on an oscilloscope-type of display as “apparent charge”, or the equivalent charge injection that would create the voltage disturbance seen. Apparent charge is measured in picocoulombs, so this is a very sensitive test method. The advantage of PD testing is that the occurrence rate of PD events can be monitored as the test voltage is increased and a potential isolation failure can be predicted before it occurs, so the test can be stopped before the converter is irretrievably damaged.

The results of the PD test need to be interpreted carefully as there may be many spurious readings before a valid result can be obtained. Therefore, a "settling" period is required to allow the charges to equalise and readings should only be taken during the last 10seconds of the test (see Fig.12). A revised version of the test protocol lasting only 1second and testing with a voltage of 1.875 × V_RATED(RMS) can be used for production testing.



Isolation Resistance and Capacitance

The input and output resistance and capacitance has to be measured with an AC signal. The isolation resistance is usually measured with 500Vdc using a Megger or similar instrument and is typically 10GΩ or higher. The isolation capacitance must be measured at a high frequency of 1MHz to eliminate the influence of on-board filter capacitors. The insolation capacitance can vary from 5pF up to 1500pF depending on the transformer construction. As with all very low current measurements, the results can be strongly affected by air humidity and temperature.

Dynamic Load Response

The Dynamic Load Response (DLR) specifies the reaction of a DC/DC converter to a step change in the load. It can be defined in two ways, by the time taken for the output voltage to return within the specified tolerance band of the output voltage or the maximum deviation of the output voltage from the nominal output voltage or a full definition of the DLR, both need to be known, but most datasheets only specify the settling time. Furthermore, some manufacturers use from 25% to 100% load, some from 50% to 100% load and some just say “25% step change” without specifying the margins, so a direct comparison between different manufacturer’s datasheets is not possible. Often, the only way to make sure is to test the converter yourself.

All converters will overshoot if the load is suddenly reduced and undershoot if the load is suddenly increased. The settling time (the largest of T_OVERSHOOT or T_UNDERSHOOT) is dependent mainly on the compensation network in the PWM controller. The network has to strike a balance between a quick reaction to a step change and not over-reacting to generate a ringing output. The ideal response is “aperiodic”, which means that the value of the output voltage deflects once in one direction only.

The top trace is aperiodic and the lower trace shows output ringing due to a poorly damped comensation network.



Many electronic loads have a step change function built-in to automatically switch between two presettable loads, but if you do not have access to an electronic load, Fig. 3.14 shows how a dynamic load test setup can be simply assembled from two load resistors and a FET driven by a square wave generator.



There are some applications where a very stable output voltage and a fast reaction to a step change in load without any output voltage ringing are both required. For example, many digital circuits have rapid changes in load but the output voltage must remain tightly regulated. If the load change can be predicted in advance or detected, it is possible to switch the compensation network from “slow” to “fast” during the load change.

With analogue controllers, this is not easy, but with a digital controller, the DLR can be made programmable. This ability is one of the biggest advantages of digital control over analogue controllers.

Just as the output voltage will change if the load suddenly changes, so the output voltage will change if the input voltage suddenly changes. This parameter is rarely defined in the datasheets as an abrupt change in the input voltage occurs only in few applications. If required, a test setup can be relatively easily built if the bench power supply has an external control input or a tracking input that can be connected to a square wave generator.

Output Ripple/Noise

All DC/DC Converters have some element of output ripple and noise. The ripple component arises from the charging and discharging currents in the output filter capacitance and has a typical frequency of either the operating frequency or double the operating frequency, depending on the topology.



Superimposed on the basic ripple wave form are switching spikes (noise) that arise from all of the parasitic effects each time the main switches change state. These occur at every peak and trough of the ripple waveform. The frequencies of the switching transients are typically orders of magnitude higher than the ripple frequency, somewhere in the MHz regions. The combined waveform gives the output Ripple/Noise (R/N) figure, which is typically measured as millivolts peak-to-peak (mVp-p). NOTE: refer to section 4.3 for the correct measurement technique.

Also superimposed on the output R/N waveform is a much slower oscillation which arises from the output voltage regulation circuit. With constant load and input voltage, the output voltage will meander up and down within the tolerance band with a frequency in the single figure Hz range. This “hunting” effect is due to hysteresis in the regulation circuit and is usually ignored in the R/N figure quoted in the datasheets as it is part of the output voltage accuracy specification and therefore not included in the R/N specification.



It is tempting to try to reduce output ripple by adding output capacitors, but although this can reduce the p-p figure slightly, it is almost impossible to filter out the ripple waveform completely. In fact, in converters that use cycle-by-cycle control, some output ripple is essential for proper regulation.

A more effective method of R/N reduction is to post-regulate the output with a linear regulator. The Power Supply Ripple Rejection (PSRR) ratio of a linear regulator is very high (up to 70dB) which makes it a very effective ripple filter.

Understanding Thermal Parameters

Introduction

Since thermal design plays such a key role in optimizing system performance, a proper assessment of the thermal performance parameters is crucial to select the most suitable DC/DC converter. The technical datasheet should not just specify the ambient operating temperature limits, but also the thermal derating, the internal power dissipation, the maximum case temperature and the thermal impedance to be able to characterise a converter’s thermal performance fully. The power dissipation can be calculated from the efficiency, but if the thermal impedance is missing from the datasheet or needs to be accurately known under the actual operating conditions of the application, then it must be derived from thermal chamber tests.

Even in the controlled environment of a thermal chamber, getting reliable measurements of the thermal behavior of modular DC/DC converters requires very careful measurement techniques. Even very low air flows distort the measurement results significantly, so the Device under Test (DUT) should be placed inside a cardboard box inside the chamber to avoid draughts from the chamber air circulation fan. The ambient temperature inside the box should be measured with a calibrated sensor positioned so that the heat generated from the converter does not directly affect the reading. The case temperature of the converter should be measured at the hottest spot (TC,MAX) as defined by the manufacturer or as discovered by thermal camera images. For very small sized converters, the attachment of a temperature sensor can itself also affect the results by conducting additional heat away from the converter along the sensor wires, so a thermocouple with as small as possible point contact should be used.

For low power converters, it can be particularly difficult to get reliable thermal impedance measurement results as the converter self-warming is not a significant source of heat. In most cases, the operating temperature range is then determined by the temperature limits of the internal components. This can be measured by fixing thermocouples to the critical components to measure the increase in temperature above ambient and then calculating the safe limits by extrapolating several readings done at 10°C ambient temperature steps. For encapsulated converters, the thermocouples must be affixed before potting to get accurate readings.

For higher power converters, the thermal impedance can be determined by the measuring the temperature rise under natural convection (still air flow) and calculating the internal power dissipation. The thermal impedance figure can then be used to also calculate the heat transfer coefficients for different rates of forced convection.

Finally, very low temperatures also adversely affect converter performance. The lower temperature limit depends on one of three factors, whichever arises first: the minimum temperature rating of the components used, a reduced gain or shift in the bias point of the PWM circuit which prevents the converter from starting up, or mechanical failure caused by mismatched thermal contraction coefficients.

Thermal Impedance

To begin the analysis of DC/DC thermal performance, let us consider thermal impedance.

Thermal impedance or thermal Resistance is a measure of how effective the heat transfer is between an internal heat source such as the transformer core or semiconductor junction and the surroundings. Consider the switching FET, for example. The heat source is the semiconductor junction, T_J. The heat is transferred to the transistor body (T_B), then through the potting medium to the converter case (T_C) and finally from the case to the surroundings (T_AMB). Each of these stages will have a thermal impedance, θ, measured in °C/W or thermal resistance, RTH, measured in °K/W.

The two are for all practical purposes completely interchangeable.



Of the thermal impedances mentioned above, the user can influence only the last impedance in the chain, θ_CA, as the other two impedances are fixed as part of the thermal management design of the converter.

The temperature rise of the converter can be calculated using:



Calculation Example: the RECOM RP15-4805SA converter has an output power of 15W, an efficiency of 88% and a case-ambient thermal impedance of 18.2°C/W. The maximum allowed case temperature is 105°C. The power dissipation = 15/0.88 - 15 = 2.04W and the associated temperature rise of the case above ambient = 37°C. The maximum allowable ambient temperature is thus = 105°C - 37°C = 68°C.

If the thermal impedance is not known, it can be derived by measurement. For an approximate value, a thermal chamber is not necessary. A suitable test set up is shown in Fig. 3.18. As with all thermal measurements, sufficient time should be left to allow the temperatures to settle before taking any readings.



The thermal impedance can be derived from rearranging Equation 3.8:



As the power dissipation is known (the difference between the input power and the output power), then measuring the internal case temperature rise above ambient allows the thermal impedance to be determined.

Thermal Derating

All DC/DC converters dissipate power internally as heat and so run warmer than their surroundings. As long as this additional warmth can be transferred away to the surroundings, the converter can run at full power. However, as the ambient temperature increases, it becomes increasingly difficult to lose this excess heat. At a certain ambient temperature, the converter reaches its maximum temperature limit and any further increases in ambient temperature must be compensated for by reducing the amount of power dissipated inside the converter by reducing the load. This is called thermal derating.

Fig. 3.19 shows an example derating curve, again for the RP15-4805SA converter used in the previous example. The converter can be used at full power over the ambient temperature range of -40°C to +68°C. If the application has a specification that the maximum ambient temperature must be up to 85°C, then the converter load must be reduced to 55% to allow operation from -40°C to +85°C.



There is a practical limit to the amount of derating possible. The derating curve assumes that the efficiency remains stable as the load decreases, but this is not true at very low loads. In practice, derating much below 40% load is counter-productive as the reduction in power dissipation due to load reduction is negated by the increase in power dissipation due to the lower efficiency. Fig. 3.20 shows how the power dissipation curve flattens out and may even start to increase again at low load.



Adding a heatsink to a converter improves the heat transfer to the surroundings (reduces θ_CA) and thus extends the maximum operating temperature range. However, despite its name, heatsinks do not absorb heat. The heat transferred to the heatsink must still eventually be transferred to the surroundings. Its function is therefore to increase the effective surface area of the converter. The RP15-4805SA used in this example is a very compact converter with a 1” × 1” case size. Thus the heatsink that fits onto the converter is also very compact and therefore does not increase the surface area dramatically. In general, clip on heatsinks bring only 5 - 10°C increase in the maximum operating temperature range, unless it is itself cooled by other methods such as forced convection.



Furthermore, heatsinks can only be used with metal-cased or baseplate-mounted converters. Adding a heatsink to a plastic cased converter is actually counter-productive, because the heatsink has a poor thermal connection to the plastic and blocks the convection airflow.

In conclusion, derating is useful to get that extra 10 - 15°C extension to the upper operating temperature range, but that is about the limit for many applications. Heatsinking can help, but if the heatsink is similarly dimensioned as the converter, then again only an extra 5 - 15°C extension to the upper operating temperature range can be expected.

To extend the the maximum temperature range significantly requires forced cooling.

Forced Cooling

Forced convection cooling (fan cooling) reduces the thermal impedance to ambient, θ_CA by adding heat advection to the natural convection. Advection is dependent on the mass of air flowing past the converter per second and also to the turbulence of the air flow. If natural convection is given a normalised thermal impedance of 1.00, then with increasing laminar air flow (given in Linear Feet per Minute, LFM), the thermal impedance reduces thus:



If we return to our example converter used to derive the Fig. 3.19 derating graph. The temperature rise equation is still valid: ΔT_RISE = PDISS θ_CA and the internal power dissipation is still the same, 2W. With natural convection, the θCA value was 18.2°C/W giving a temperature rise of 37°C, leading to a maximum full load operating temperature of 68°C. With 100LFM forced convection, the θCA value would by multiplied by 0.67 or be 12.2°C/W giving a reduced temperature rise of 24.4°C, leading to a maximum full load operating temperature of 81°C.

The combination of forced convection and a small heatsink is slightly more effective than forced convection alone, but we are getting into the law of diminishing returns. Adding a clip-on heatsink increased the maximum full load operating temperature from 68°C to 73°C with natural convection, an increase of 5°C. Calculating backwards, we can determine that the θCA dropped from 18.2°C/W without the heatsink to around 16°C/W with the heatsink. With 100LFM, this thermal impedance would reduce further to 10.5W/°C. This would increase the maximum full load operating temperature to 84°C, or only 3°C higher than 100LFM without a heatsink. Fig. 3.22 shows the calculated results for the converter with the heatsink fitted at different air flows:



Conducted and Radiated Cooling

There are other transport mechanisms for thermal transfer than convection or advection. Heat can also be removed from a converter by conduction and radiation.

Thermal conduction is the transfer of heat from one object to another across a thermal gradient via direct contact. The transport mechanism is phonons, or transfer of energy from one molecule to another by atomic level vibration. The transfer rate, or thermal flux, is dependent on the temperature difference and the thermal conductivity of the material (measured in Wm-1°C-1).

Converters fitted with baseplates rely on thermal conduction as the primary cooling method, but all converters can benefit from some conducted heat transfer via the connection pins into the PCB tracks. The thermal impedance chain for a baseplate- cooled converter is shown in Figure 3.23:



As the primary heat flux transfer is by direct contact, it is very important that the contact area between the baseplate and heatsink is as large as possible. Even very small imperfections will leave air gaps across which there will be no conducted heat transfer. If both the baseplate and heatsink are not flat to within 5mils (0.125mm) then the thermal conduction will be sharply impaired. Therefore it is common to use a Thermal Interface Medium (TIM) such as thermal grease or a gap pad to ensure a good physical and thermal contact.

Radiated heat transfer is the transfer of heat from a body by infra-red radiation. The heat of the sun that we feel is purely radiated heat as the vacuum of space blocks all convected and conducted heat transfer. A converter can also dissipate energy in a vacuum by radiated heat loss, but the low converter case temperature of around 300 - 400°K makes this form of heat loss very small in comparison to conducted or convected heat transfer. At high altitudes, however, cooling by both conduction and radiation become increasingly important as the main disadvantage of convected heat transfer is that it is dependent on the air flow mass and thus decreases as the air pressure drops.